Quadrature transmitter, wireless communication unit, and method for spur suppression

ABSTRACT

A quadrature transmitter includes a first and second matched transmitter path. Each transmitter path receives respective sets of quadrature baseband signals. At least one local oscillator port receives respective sets of quadrature LO signals. Mixer stage(s) respectively multiply the sets of quadrature baseband signals with the respective sets of quadrature LO signals to produce a respective output radio frequency signal. A combiner combines the output RF signals. The first set of quadrature signals is a substantially 45° phase shifted version of the second set of quadrature signals; and the first set of quadrature LO signals is a reverse substantially 45° phase shifted version of the second set of quadrature LO signals. A baseband error correction circuit corrects a phase error between the quadrature baseband signals at baseband and a LO error correction circuit corrects a phase error between the quadrature baseband signals at a LO frequency.

FIELD OF THE INVENTION

The field of this invention relates generally to the field oftransmitters in wireless communication units. In particular, the fieldof this invention relates to suppression of both counter intermodulationproducts and harmonic spurs.

BACKGROUND OF THE INVENTION

As new generations of handsets, and other wireless communication devicesbecome embedded with more applications and complexity, there is a needfor ever more integration. The trend in mobile radio communications istowards complex multi-radio systems comprised of several paralleltransceivers. This implies a leap in complexity of the radio frequency(RF) front-end (FE) design. The RF circuits of wireless communicationdevices, and the transmitter parts in particular, are difficult tointegrate.

Known transmitter architectures create undesired harmonics of transmitsignals, due to the nonlinearity of each transmitter stage, e.g. ananalog quadrature baseband circuit, an up-conversion mixer stage, thepower amplifier (PA) stage, etc. This results in harmonic RF spurs beinggenerated at the transmit output, which may not comply with out-of-bandtransmission specifications of wireless communication standards and thusimpact the communication transmission and reception of other wirelesscommunication units. Alternatively, or additionally, they may causeself-interference in other transceiver paths implemented in the samecommunication unit.

In particular, spurs may be generated around the frequency of thewanted/desired transmit signal, ω₀+ω_(bb), that are within the transmitband. Such undesired spurs and harmonics include the baseband imagefrequency at the RF, ω₀−ω_(bb), the local oscillator leakage, ω₀ LOleakage, and multiple counter inter-modulation products (referred to asCIM spurs), such as third, fifth, seventh, . . . harmonics of thebaseband signal located around the wanted/desired transmit signal,ω₀−3ω_(bb), ω₀+5ω_(bb). In particular, CIM spurs around the wantedsignal severely degrade performance such as adjacent channel leakagerejection (ACLR) and spurious emissions. Counter 3^(rd) order and 5^(th)order intermodulation (CIM3/CIM5) components are known to be the mostcritical ones to cancel or remove, with the higher order CIM productsbeing less significant.

Harmonic mixing at different TX stages generates/regenerates CIMproducts. As the harmonics are regenerated at each TX stage, it isnecessary to suppress CIM products. Notably, due to the problematiceffect that each active stage regenerates CIM products, even if theyhave been substantially cancelled or removed earlier in the transmitchain, all the stages of the transmitter need to be considered whenattempting to remove the generated/regenerated CIM products.

Four known solutions of: Weldon, He, Vora and Ingels attempt to reduceharmonic spurs, each of which are based on essentially the same idea ofsinewave approximation. By adding multiple signals with different phasesand amplitudes, a first order sinewave approximation can be achieved. Anamplitude scaling of √2 allows the use of phase shifts that are easy togenerate and hence each of the known art that is identified below use √2amplitude scaling in the signal path. One disadvantage common to thesefour solutions is that RF signals with different phases are combineddirectly, inevitably leading to power loss and hence reduction oftransmitter efficiency.

FIG. 1 illustrates a known quadrature transmitter architecture 100. Thetransmitter architecture 100 comprises a quadrature (I/Q) baseband inputsignal 110. The I/Q baseband input signal 110 is input to quadratureup-mixer 130 via a respective low pass filter 120, which up-converts theI/Q baseband signal 110 in response to respective quadrature localoscillator (LO) signals 125, 135, there being a 90 degree phase shiftbetween the respective quadrature LO signals. The up-convertedquadrature signals are amplified in RF amplifiers 140 and both paths aresummed at combiner 150. The combined signal is then amplified in poweramplifier 170.

FIG. 2 provides a graphical illustration 200 of a wanted signal 210 anda multitude of spurs and harmonics that are created in known quadraturetransmitters, that require careful attention when designing atransmitter architecture. The illustrated spurs and harmonics include LOleakage 226 (ω₀), image spur 224, CIM3 spur 220 (ω₀−3ω_(bb)) and CIM5spur 232 (ω₀+5ω_(bb)), as well as other 2^(nd) harmonic spurs 222 and228 and 3^(rd) harmonic spur 230.

A transmitter architecture by M. Ingels et al., as described in “Amultiband 40 nm CMOS LTE SAW-less modulator with −60 dBc C-IM3”,published in the ISSCC digest of technical papers, p338-339, February2013, targets rejection/cancellation of only the C-IM3 product. Forexample, in Ingels, the CIM3 products from the three paths have threephases, 0, −90°, 135°. With a scaling factor of √2, the CIM3 tone iscancelled when the three signals are summed. Thus, Ingels (as well asother known techniques) describes an architecture that is capable ofcareful manipulation of quadrature signals to effect cancellation of oneundesired transmit harmonic or spur, unfortunately leaving other strongharmonics and spurs that fail to meet a specific performance or mayre-create undesired spurious emissions.

Therefore, known techniques for reducing or cancelling harmonic spurs,including CIM products, are less than ideal in that other harmonicspurs, including CIM products, are generated at sufficient levels tocause or potentially cause spurious emission issues.

SUMMARY OF THE INVENTION

Accordingly, the invention seeks to mitigate, alleviate or eliminate oneor more of the above mentioned disadvantages, either singly or in anycombination. Aspects of the invention provide a quadrature transmitter,a wireless communication unit and a method of spur reduction orcancellation, as described in the appended claims.

In accordance with a first aspect of the invention, a quadraturetransmitter comprises: a first transmitter path and a second transmitterpath that are matched. Each transmitter path comprises: at least oneinput arranged to receive respective first or second sets of quadraturebaseband signals; at least one local oscillator, LO, port configured toreceive respective first and second sets of quadrature LO signals; andat least one mixer stage coupled to the at least one input andconfigured to respectively multiply the sets of quadrature basebandsignals with the respective first or second sets of quadrature LOsignals to produce a respective output radio frequency, RF, signal. Acombiner is configured to combine the output radio frequency signals ofthe first transmitter path and the second transmitter path. The firstset of quadrature baseband signals is a substantially 45° phase shiftedversion of the second set of quadrature baseband signals; and the firstset of quadrature LO signals is a reverse substantially 45° phaseshifted version of the second set of quadrature LO signals.

In this manner, by implementing the aforementioned architecture, it ispossible to reduce or cancel multiple problematic harmonic spurs, aswell as CIM products.

In some optional examples, at least two sets of undesired radiofrequency signals are anti-phase such that they are cancelled in thecombiner.

In some optional examples, a single set of quadrature baseband signalsis applied to both the first transmitter path and the second transmitterpath, where only one of the first and second transmitter paths maycomprise at least one baseband phase shifter coupled to the quadratureinput and configured to provide a phase shifted by substantially ±45°representation of the input signal of the set of quadrature signalsapplied in the other transmitter path to its respective baseband input.In some optional examples, the baseband phase shift may be performed ina digital domain and the first and second transmitter path each comprisea set of digital to analog converters, DACs, configured to generateanalog quadrature signals.

In some optional examples, the quadrature transmitter may furthercomprise at least one error correction circuit operably coupled to aninput of a set of DACs configured to correct non-idealities in the setof quadrature analog signals between the first and second transmitterpaths.

In some optional examples, each of the first transmitter path and secondtransmitter path comprises at least one error correction circuitoperably coupled to an input of the at least one DAC, wherein the atleast two error correction circuits may be configured to correctnon-idealities on a respective first transmitter path or secondtransmitter path separately in a digital domain before or after a phaserotation of a set of quadrature analog signals in the digital domain.

In some optional examples, the quadrature transmitter may furthercomprise three error correction circuits operably coupled to an input ofthe at least one DAC wherein two error correction circuits areconfigured to correct mismatches within a respective first and secondtransmitter path and a third error correction circuit is configured tocorrect mismatches on both the first and second transmitter paths.

In some optional examples, one of the transmitter paths may includeanalog phase rotation where the single set of quadrature basebandsignals is converted by a set of digital to analog converters, DACs,configured to generate analog quadrature signals. In some optionalexamples, at least one error correction circuit operably coupled to aninput of the set of DACs configured to apply one or more corrections tothe set of quadrature digital signals shared between the first andsecond transmitter paths

In some optional examples, the quadrature transmitter may furthercomprise a controller coupled to first pairs of switches coupled to thebaseband input of the second transmit path and second pairs of switcheslocated on the LO path of the second transmitter path and configured toselectively reconfigure the second transmitter path to operate on thesame set of LO and baseband signals as the first transmitter path.

In some optional examples, each of the first transmitter path and secondtransmitter path may be implemented as a plurality of sliced transmitterpaths where the combiner is a power combiner located external to thefirst plurality of sliced RF modules and second plurality of sliced RFmodules. In some optional examples, each of the first plurality ofsliced RF modules and second plurality of sliced RF modules may comprisea second combiner configured to combine RF quadrature signals output byrespective sliced RF modules.

In some optional examples, the quadrature transmitter may furthercomprise a controller coupled to a first pair of switches coupled to thebaseband input of the second transmit path to provide the first set ofquadrature baseband signals to each sliced RF module of the secondtransmitter path; and second pairs of switches located on the LO path ofeach respective sliced RF module of the second transmitter path andconfigured to selectively apply the reverse phase shifts to mixer stageson each sliced RF module of the second transmitter path.

In accordance with a second aspect of the invention, a communicationunit comprising a quadrature transmitter that comprises: a firsttransmitter path and a second transmitter path that are matched andwherein each transmitter path comprises: at least one input arranged toreceive respective first or second sets of quadrature baseband signals;at least one local oscillator, LO, port configured to receive respectivefirst and second sets of quadrature LO signals; and at least one mixerstage coupled to the at least one input and configured to respectivelymultiply the sets of quadrature baseband signals with the respectivefirst or second sets of quadrature LO signals to produce a respectiveoutput radio frequency, RF, signal; and a combiner configured to combinethe output radio frequency signals of the first transmitter path and thesecond transmitter path. The first set of quadrature baseband signals isa substantially 45° phase shifted version of the second set ofquadrature baseband signals; and the first set of quadrature LO signalsis a reverse substantially 45° phase shifted version of the second setof quadrature LO signals.

In accordance with a third aspect of the invention, a method for atransmitter that comprises a first transmitter path and a secondtransmitter path that are matched, the method comprising: receiving afirst set of quadrature baseband signals at the first transmitter path;receiving a second set of quadrature baseband signals at the secondtransmitter path; generating respective first and second sets ofquadrature LO signals; multiplying the first and second sets ofquadrature baseband signals with the respective first or second sets ofquadrature LO signals to produce a respective output radio frequency,RF, signal; and combining the output radio frequency signals of thefirst transmitter path and the second transmitter path. The first set ofquadrature baseband signals is a substantially 45° phase shifted versionof the second set of quadrature baseband signals; and the first set ofquadrature LO signals is a reverse substantially 45° phase shiftedversion of the second set of quadrature LO signals.

These and other aspects of the invention will be apparent from, andelucidated with reference to, the embodiments described hereinafter.

BRIEF DESCRIPTION OF THE DRAWINGS

Further details, aspects and embodiments of the invention will bedescribed, by way of example only, with reference to the drawings.Elements in the figures are illustrated for simplicity and clarity andhave not necessarily been drawn to scale. Like reference numerals havebeen included in the respective drawings to ease understanding.

FIG. 1 illustrates a known transmitter architecture, particularly someof the components that create undesired harmonics of transmit signals.

FIG. 2 illustrates a frequency response graph of a known transmitterarchitecture that highlights CIM products and harmonics and spurs thatrequire cancelling or reducing.

FIG. 3 illustrates a wireless communication unit comprising atransmitter architecture adapted according to a second exampleembodiment of the invention.

FIG. 4 illustrates an example overview of a transmitter architectureaccording to example embodiments of the invention.

FIG. 5 illustrates an example overview of an alternative transmitterarchitecture embodiment using opposite polarity rotation on thesecondary signal path on both the BB and LO, according to exampleembodiments of the invention.

FIG. 6 illustrates an example overview of a yet further alternativetransmitter architecture using one pair of DACs and analog BB phaserotation according to example embodiments of the invention.

FIG. 7 illustrates an example overview of a still yet furtheralternative embodiment using two pairs of DACs and digital BB phaserotation (with optional error correction on each transmitter path),according to example embodiments of the invention.

FIG. 8 illustrates an example overview of a still yet furtheralternative embodiment where a sliced transmitter can be selectivelyreconfigured to perform the harmonic/spur rejection or cancellationaccording to example embodiments of the invention.

FIG. 9 illustrates a graphical example of a simulation of phasesensitivity highlighting the attenuation applied to respective multipleharmonics or spurs, according to example embodiments of the invention.

FIG. 10 illustrates a graphical example of a simulation of amplitudesensitivity highlighting the attenuation applied to respective multipleharmonics or spurs, according to example embodiments of the invention.

FIG. 11 illustrates an example flowchart to cancel or reject multipleharmonics or spurs, according to example embodiments of the invention.

FIG. 12 illustrates an example overview of a still yet furtheralternative embodiment where a sliced transmitter can be selectivelyreconfigured to perform the harmonic/spur rejection or cancellation withboth baseband phase rotation and local oscillator phase rotation,according to example embodiments of the invention.

FIG. 13 illustrates an example flowchart to cancel or reject multipleharmonics or spurs with both phase rotation and error correction atbaseband to correct baseband phase errors and local oscillator phaserotation and error correction at an LO frequency to correct LO phaseerrors, according to example embodiments of the invention.

DETAILED DESCRIPTION

Examples of the invention will be described in terms of directconversion quadrature transmitters with at least two matched quadraturepaths. Although most examples of the invention will be described interms of a single-ended implementation, as shown in FIG. 4, it isenvisaged that the concepts described herein may be equally applied toany differential or double-balanced implementation.

The inventors have recognised and appreciated that although thenon-linearity of each block in FIG. 1 is an important figure of merit(FOM) for a transmitter design, ultimately what matters is thenonlinearity of the whole transmitter. Thus, when the respectivequadrature signals are combined, undesired spurs, harmonics and CIMproducts should be cancelled at this point.

Examples of the invention will be described in terms of using two setsof quadrature (I/Q) signals with 45° phase offset for both baseband andLO. Notably, none of the examples of the invention employ any √2 signalpath scaling. Further, no power loss of the transmit signal is incurred,since the desired signal from both (or multiple) quadrature paths iscombined in-phase.

This invention can be applied to any transmitter path architecture.Furthermore, the spurs described here will be cancelled but depending onthe architecture other harmonic spurs may exist that are not cancelled.The mixers described throughout this description can be advantageouslyimplemented using any known mixer architecture, including active orpassive mixers and voltage mode or current mode mixers.

Advantageously, with the various transmit architectures describedherein, multiple problematic harmonics and CIM products that aregenerated or re-generated are cancelled, irrespective of where in thetransmitter path the harmonics or CIM products occur.

Referring now to FIG. 3, a block diagram of a wireless communicationunit 300, adapted in accordance with some example embodiments of theinvention, is shown. In practice, purely for the purposes of explainingembodiments of the invention, the wireless communication unit isdescribed in terms of a wireless subscriber communication unit, which insome examples may be a smartphone.

The wireless communication unit 300 contains an antenna arrangement 302,for radiating signals and/or for receiving transmissions, coupled to anantenna switch or duplexer 304 that provides isolation between receiveand transmit chains within the wireless communication unit 300.

One or more receiver chains, as known in the art, include(s) receiverfront-end circuitry 306 (effectively providing reception, filtering andintermediate or base-band frequency conversion). The receiver front-endcircuitry 306 is coupled to a signal processing module 308 (generallyrealized by a Digital Signal Processor (DSP)). A skilled artisan willappreciate that the level of integration of receiver circuits orcomponents may be, in some instances, implementation-dependent.

A controller 314 maintains overall operational control of the wirelesscommunication unit 300. The controller 314 is coupled to the receiverfront-end circuitry 306 and the signal processing module 308. In someexamples, the controller 314 is also coupled to a buffer module 317 anda memory device 316 that selectively stores operating regimes, such asinformation relating to quadrature phase and amplitude settings to beapplied to components in the transmitter to suppress harmonic spurs andCIM products, and the like. A timer 318 is operably coupled to thecontroller 314 to control the timing of operations (e.g. transmission orreception of time-dependent signals) within the wireless communicationunit 300.

The transmit chain includes transmitter/modulation circuitry 322 and apower amplifier 324 coupled to the antenna arrangement 302, which maycomprise for example an antenna array, or plurality of antennas. Thetransmitter/modulation circuitry 322 and the power amplifier 324 areoperationally responsive to the controller 314. In some examples, thesignal processing module 308 and/or controller 314 may receive inputsfrom one or more input device or sensor modules 320.

Frequency generation circuit 328 is operably coupled to the receiverfront-end circuitry 306 and the transmitter/modulation circuitry 322and, in accordance with example embodiments, is arranged to providequadrature local oscillator signals 329 of various phases thereto. Inexample embodiments, the transmit chain is a quadrature transmitter thatcomprises a first transmitter path and a second transmitter path thatare matched. Each transmitter path comprises: at least one inputarranged to receive respective first or second sets of quadraturebaseband signals 340. Frequency generation circuit 328 comprises; atleast one local oscillator, LO, 327 configured to generate respectivefirst and second sets of quadrature LO signals 329; and at least onemixer stage 321 coupled to the at least one quadrature baseband inputand configured to respectively multiply the sets of quadrature basebandsignals 340 with the respective first or second sets of quadrature LOsignals 329 to produce a respective output radio frequency, RF,signal(s) 350. A combiner is configured to combine the output radiofrequency signals of the first transmitter path and the secondtransmitter path and may be located within the transmitter/modulationcircuitry 322, or within a module supporting the power amplifier 324 orsomewhere there between. Notably, in accordance with exampleembodiments, a first set of quadrature baseband signals is a 45° phaseshifted version of the second set of quadrature baseband signals; andthe first set of quadrature LO signals is a reverse 45° phase shiftedversion of the second set of quadrature LO signals.

In the context of the examples herein described harmonics and CIMproducts are arranged to be cancelled by combining the harmonics andproducts with corresponding out-of-phase harmonics and products. Asunderstood to a skilled artisan, for example in the context of mixers,harmonics are generated in each mixer path, and these harmonics willhave different phases as a result of a different BB phase and LO phase.The general concept of harmonic cancellation may be achieved eventually,when all the paths are summed at a combiner. At the summation point, ifall undesired harmonics are summed to zero (i.e. the phase relationshipcancels out the harmonics) harmonic cancellation is achieved.

In accordance with examples of the invention, a first set of quadraturebaseband signals is configured to be a 45° phase shifted version of asecond set of quadrature baseband signals; and a first set of quadratureLO signals is configured to be a reverse 45° phase shifted version of asecond set of quadrature LO signals, such that when they are combinedeach of the respective undesired harmonics and CIM products aresubstantially cancelled.

Clearly, a number of the various components within the wirelesscommunication unit 300 can be realized in discrete or integratedcomponent form, with an ultimate structure therefore beingapplication-specific or design-based.

Referring now to FIG. 4, an example of a transmitter architecture 400adapted according to embodiments of the invention is described. It isenvisaged that, in some examples, the two transmitter paths in FIG. 4may include at least some part of conventional quadrature transmitters,whose signals passing there through have been adapted by careful andselective control of phase shifts.

The transmitter architecture 400 employs a +45 degree (e.g. π/4) phaseshift at the LO, and a −45 degree (−π/4) phase shift at baseband (BB)(or vice versa as will be described later with reference to FIG. 5). Inthe example embodiments, CIM3 and CIM5 are advantageously cancelled, aswell as 3^(rd) order harmonics, which would regenerate the CM productsin the subsequent PA stages.

The transmitter architecture 400 comprises a first quadrature (I/Q)baseband input signal 431. The first I/Q baseband input signal 431includes a first quadrature signal I=A(t). cos {φ(t)}) input to firstquadrature up-mixer 434 and second quadrature signal (Q=A(t). sin{φ(t)}) input to second quadrature up-mixer 435, which up-convert thefirst I/Q baseband signals 431 in response to respective quadraturelocal oscillator (LO) signals 471, 472. There is a 90 degree phase shiftbetween first quadrature LO signal 471 and second quadrature LO signal472, as well as a 90 degree phase shift between the first quadrature ‘I’signal and second quadrature ‘Q’ signal. Each of the up converted RFsignals from mixers 434 and 435 are summed within the first quadraturetransmitter path at 454.

In accordance with example embodiments of the invention, the transmitterarchitecture 400 further comprises a second quadrature (I/Q) basebandinput signal 432 that includes a third quadrature signal (I′=A(t). cos{φ(t)−45}) that is a 45 degree phase shifted version of the firstquadrature signal I=A(t). cos {φ(t)}) of the first I/Q baseband inputsignal 431. The second quadrature (I/Q) baseband input signal 432 alsoincludes a fourth quadrature signal (Q′=A(t). sin {φ(t)−45}) that is a45 degree phase shifted version of the second quadrature signal (Q=A(t).sin {φ(t)}) of the first I/Q baseband input signal 431.

The second I/Q baseband input signal 432 is input to respective thirdquadrature mixer 436 and fourth quadrature mixer 437. In accordance withexample embodiments of the invention, third quadrature mixer 436receives the third quadrature signal (I′=A(t). cos {φ(t)−45}) and thethird quadrature local oscillator signal 473, which exerts a 45 degreephase shift to third quadrature signal (I′=A(t). cos {φ(t)−45}),compared to the phase shift applied by the first quadrature localoscillator signal 471 to the first quadrature signal. In this manner,the third quadrature signal (I′=A(t). cos {φ(t)−45}) is up-converted toa radio frequency (RF) signal by third quadrature local oscillatorsignal 473. In accordance with example embodiments of the invention,fourth quadrature mixer 437 receives the fourth quadrature signal(Q′=A(t). sin {φ(t)−45}) and the fourth quadrature local oscillatorsignal 474, which exerts a 45 degree phase shift to fourth quadraturesignal (Q′=A(t). sin {φ(t)−45}), compared to the phase shift applied bythe second quadrature local oscillator signal 472 to the secondquadrature signal. In this manner, the fourth quadrature signal(Q′=A(t). sin {φ(t)−45}) is up-converted to a radio frequency (RF)signal by fourth quadrature local oscillator signal 474. Each of the upconverted RF signals from mixers 436 and 437 are summed within the firstquadrature transmitter path at 455.

Thereafter, signals on both first and second transmitter paths aresummed at summing junction 451.

In some optional examples, the quadrature transmitter may furthercomprise a matched radio frequency amplifier circuit that includes atleast two sets of matched radio frequency, RF, amplifiers, 456 each setof matched RF amplifiers 456 being connected to a respective firsttransmitter path or second transmitter path. In some optional examples,the quadrature transmitter may further comprise a matched filter circuitthat includes at least two sets of matched filters 444, each set ofmatched filters 444 being connected to a respective first transmitterpath or second transmitter path. In some optional examples, thequadrature transmitter may further comprise a matched radio frequencyamplifier circuit that includes a set of matched radio frequency, RF,amplifiers, 466 located after each transmitter path quadrature signalhas been combined, and before the two transmitter paths are combined atsumming junction 451.

In the various architectures described herein, the nonlinearity of eachtransmitter function may be modelled as follows, where, with referenceto FIG. 4, V_(ABB) describes the quadrature baseband signals afterbaseband filters 444, V_(mix) describes the output signals of mixers434, 435, 436 or 437 and V_(PAD) describes the RF signals at the outputof amplifiers 466:

V _(A) =a ₀ +a ₁ V _(i) +a ₂ V _(i) ² +a ₃ V _(i) ³+ . . .  [1]

V _(m)=(b ₀ +b ₁ V _(A)+₂ V _(A) ² +b ₃ V _(A) ³+ . . . )(Σα_(n)cos(n)+Σβ_(n) sin(n))  [2]

V _(p) c ₀ +c ₁ V _(m) +c ₂ V _(m) ² +c ₃ V _(m) ³+ . . .  [3]

It is noteworthy that, in the modelling above, a Taylor series is usedto model the nonlinearity. If memory effects need to be taken intoaccount, other techniques such as Volterra series may be used instead.However, for the purpose of the discussion herein, memory effects do notintroduce fundamental differences. Therefore for simplicity, Taylorseries will be used throughout this discussion. The LO waveform may bemodelled using Fourier series, as appreciated by skilled artisans.

From such modelling, it is known that every single harmonic componenthas multiple contributors. For example, if we take CIM3, which is atfrequency of ω_(LO)-3ω_(BB), it is noted that amongst its multiplegenerating mechanisms, it can be created as a result of ABB 3^(rd) ordernonlinearity mixing with LO fundamental tone. Similarly, CIM3 can becreated as a result of ABB 2^(nd) order nonlinearity mixing with LO2^(nd) harmonic and then mixing with the wanted signal through PA driveramplifier (PAD) 2^(nd) nonlinearity, etc. Due to the fact that there areso many mechanisms to generate CIM3, it is impractical to go throughevery single mechanism and find a cancellation solution for each ofthem. In other words, in order to suppress the CIM3 or any otherharmonic component, a solution to fix all the possible mechanisms atonce is advantageous.

One interesting property of the signal path is that, if a phase shift φis introduced at an analog quadrature baseband (ABB) input V_(i), thephase shift at the signal path output is nφ for any harmonic componentsat a frequency of mω_(LO)+nω_(BB). In the same way, a phase shift θintroduced at an LO, results in a phase shift at the signal path outputof mθ for harmonic components at a frequency of mω_(LO)+nω_(BB).

An extra identical signal path can be introduced, with the outputs beingsummed either in voltage or current. With a correct phase shiftintroduced on both the LO and ABB input of the identical path, if theoriginal signal path and the secondary signal path are substantiallymatched, then a perfect cancellation of those harmonics that arecompletely out of phase in one path compared to the other path can beachieved.

Herein, in the various illustrated architectures, we propose a π/4 phaseshift on the LO and −π/4 phase shift on the ABB input on the secondarysignal path, for example. With these phase shifts, it can be easilyshown that, for CIM3, the phase shift on the secondary path output isπ/4−(−π/4)*3=π. Similarly, for a CIM5 product (a harmonic spur atω_(LO)+5ω_(BB)), the phase shift on the secondary path output isπ/4+(−π/4)*5=−π. As both these phases are exactly out of phase comparedto the first path, CIM3 and CIM5 will be cancelled out at the summedoutput of the two matched transmitter paths.

Whilst the CIM3/CIM5 products are being cancelled, the wanted signal(ω_(LO)+ω_(BB)) at the secondary path has a phase shift of π/4+(−π/4)=0,which means the wanted signal on the two transmitter paths are in phase.Thus, there is advantageously no power loss for the wanted signal whenthe outputs of the two transmitter paths are summed.

It is not usually desirable to include the PA's into the cancellationpath. We also would like to suppress the harmonics (which themselves arenot at CIM3/CIM5 frequencies), but will regenerate CIM3/CIM5 at the PAoutput due to PA nonlinearity if not dealt with earlier. Amongst them,the critical harmonics are known to be at 3ω_(LO)−ω_(BB), whichgenerates CIM3, and at 3ω_(LO)+7ω_(BB), which generates CIM5. Thesetones are also cancelled out in a similar way as the CIM3/CIM5cancellation described above. The phase shift for the 3ω_(LO)−ω_(BB)tone in the second transmitter path is (π/4)*3−(−π/4)=π which again isout of phase compared to the first transmitter path. Similarly, thephase shift for the 3ω_(LO)+7ω_(BB) tone in the second transmitter pathis (π/4)*3+(−π/4)*7=−π. Hence, these undesired terms may also becancelled.

Thus, the techniques described herein substantially cancel out both CIM3and CIM5 products from the transmitter path output, without anysacrifice of power efficiency. Advantageously, the techniques describedherein also substantially suppress the harmonic spurs that willre-generate CIM3/CIM5 at the PA output. In this manner, a solution isprovided to substantially prevent the generation of CIM3, CIM5 and3^(rd) harmonic products for the whole transmitter system, irrespectiveof whether the undesired products are created earlier as they areultimately cancelled out.

Thus, for example in the architecture of FIG. 4 (and the othersdescribed later), it can be shown that harmonic rejection of the CIM3and CIM5 products, as well as suppression of harmonic spurs (that couldsubsequently lead to re-generation of CIM3 and CIM5 products) can beachieved in this circuit.

FIG. 5 illustrates an example overview of an alternative transmitterarchitecture embodiment using opposite polarity rotation on thesecondary signal path on both the BB and LO, according to exampleembodiments of the invention.

The transmitter architecture 500 employs a −45 degree (e.g. −π/4) phaseshift at the LO, and a +45 degree (+π/4) phase shift at baseband (BB).In the example embodiments, CIM3 and CIM5 are advantageously cancelled,as well as 3^(rd) order harmonics, which could regenerate the CIMproducts and further harmonics in the subsequent transmit chain stages,if not supressed. Thus, some examples of the inventive concept proposeboth a +45 degree phase shift at BB and a −45 degree at LO, compared tothe conventional quadrature (IQ) transmitter.

The transmitter architecture 500 comprises a first quadrature (I/Q)baseband input signal 531. The first I/Q baseband input signal 531includes a first quadrature signal (I=A(t). cos {φ(t)}) input to firstquadrature up-mixer 534 and second quadrature signal (Q=A(t). sin{φ(t)}) input to second quadrature up-mixer 535, which up-converts thefirst I/Q baseband signals 531 in response to respective quadraturelocal oscillator (LO) signals 571, 572. There is a 90 degree phase shiftbetween first quadrature LO signal 571 and second quadrature LO signal572. Each of the up converted RF signals from mixers 534 and 535 aresummed within the first quadrature transmitter path at 554.

In accordance with example embodiments of the invention, the transmitterarchitecture 500 further comprises a second quadrature (I/Q) basebandinput signal 532 that includes a third quadrature signal (I′=A(t). cos{φ(t)+45}) that is a 45 degree phase shifted version of the firstquadrature signal (I=A(t). cos {φ(t)}) of the first I/Q baseband inputsignal 531. The second quadrature (I/Q) baseband input signal 532 alsoincludes a fourth quadrature signal (Q′=A(t). sin {(φ(t)+45}) that is a45 degree phase shifted version of the second quadrature signal (Q=A(t).sin {φ(t)}) of the first I/Q baseband input signal 531.

The second I/Q baseband input signal 532 is input to respective thirdquadrature mixer 536 and fourth quadrature mixer 537. In accordance withexample embodiments of the invention, third quadrature mixer 536receives the third quadrature signal (I′=A(t). cos {φ(t)+45}) and thethird quadrature local oscillator signal 573, which exerts a −45 degreephase shift to third quadrature signal (I′=A(t). cos {φ(t)+45}),compared to the phase shift applied by the first quadrature localoscillator signal 571 to the first quadrature signal. In this manner,the third quadrature signal (I′=A(t). cos {φ(t)+45}) is up-converted toa radio frequency (RF) signal by third quadrature local oscillatorsignal 573. In accordance with example embodiments of the invention,fourth quadrature mixer 537 receives the fourth quadrature signal(Q′=A(t). sin {φ(t)+45}) and the fourth quadrature local oscillatorsignal 574, which exerts a −45 degree phase shift to fourth quadraturesignal (Q′=A(t). sin {φ(t)+45}), compared to the phase shift applied bythe second quadrature local oscillator signal 572 to the secondquadrature signal. In this manner, the fourth quadrature signal(Q′=A(t). sin {φ(t)+45}) is up-converted to a radio frequency (RF)signal by fourth quadrature local oscillator signal 574. Each of the upconverted RF signals from mixers 536 and 537 are summed within the firstquadrature transmitter path at 555.

Thereafter, signals on both first and second transmitter paths aresummed at summing junction 551. In some optional examples, thequadrature transmitter may further comprise a matched radio frequencyamplifier circuit that includes at least two sets of matched radiofrequency, RF, amplifiers, 556 each set of matched RF amplifiers 556being connected to a respective first transmitter path or secondtransmitter path. In some optional examples, the quadrature transmittermay further comprise a matched filter circuit that includes at least twosets of matched filters 544, each set of matched filters 544 beingconnected to a respective first transmitter path or second transmitterpath. In some optional examples, the quadrature transmitter may furthercomprise a matched radio frequency amplifier circuit that includes a setof matched radio frequency, RF, amplifiers, 566 located after eachtransmitter path quadrature signal has been combined, and before the twotransmitter paths are combined at summing junction 551.

FIG. 6 illustrates an example overview of a yet further alternativetransmitter architecture using one pair of DACs and analog quadraturebaseband (BB) phase rotation according to example embodiments of theinvention. In the example embodiments, CIM3 and CIM5 are againadvantageously cancelled, as well as 3^(rd) order harmonics, which couldregenerate the CIM products and further harmonics in the subsequenttransmit chain stages, if not supressed. Again, the transmitterarchitecture 600 employs a 45 degree (e.g. −π/4) phase shift at the LO,and a reverse 45 degree (π/4) phase shift at baseband (BB). In thisexample, the directions of the phase rotation are not specified, as itcan be appreciated that the BB and LO respectively apply reverse (i.e.opposite) 45 degree phase rotations.

The transmitter architecture 600 comprises a first quadrature (I/Q)baseband input signal 631. The first and second I/Q baseband inputsignals 631, 632 may be subjected to error correction in the digitaldomain in error correction circuit 620 and then converted into theanalog domain by a pair of digital-to-analog converters (DACs) 680.Thus, in essence in this example, the error correction circuit 620applies corrections to a single I/Q baseband input signal of a compositetransmit system implemented with two paths. In this example, the analogfirst I/Q baseband input signal 631 is then input to a baseband 45degree phase rotation circuit 690 that imparts a 45 degree phaserotation to the input signal to create an analog second I/Q basebandinput signal 632. In the same manner as FIG. 4 and FIG. 5, analog firstI/Q baseband input signal 631 is input to first quadrature up-mixers634, 635, which up-convert the first I/Q baseband signals 631 inresponse to respective quadrature local oscillator (LO) signals. Thereis a 90 degree phase shift between first quadrature LO signal 671 andsecond quadrature LO signal 672. Each of the up converted RF signalsfrom mixers 634 and 635 are summed within the first quadraturetransmitter path at 654.

In accordance with example embodiments of the invention, the analogsecond I/Q baseband input signal 632 includes a third quadrature signalthat is a 45 degree phase shifted version of the first quadrature signalof the analog first I/Q baseband input signal 631. The analog second I/Qbaseband input signal 632 also includes a fourth quadrature signal thatis a 45 degree phase shifted version of the second quadrature signal ofthe analog first I/Q baseband input signal 631.

The second I/Q baseband input signal 632 is input to respective thirdquadrature mixer 636 and fourth quadrature mixer 637. In accordance withexample embodiments of the invention, third quadrature mixer 636receives the third quadrature signal and the third quadrature localoscillator signal 673, which exerts a reverse 45 degree phase shift tothe third quadrature signal, compared to the phase shift applied by thefirst quadrature local oscillator signal 671 to the first quadraturesignal. In this manner, the third quadrature signal is up-converted to aradio frequency (RF) signal by third quadrature local oscillator signal673. In accordance with example embodiments of the invention, fourthquadrature mixer 637 receives the fourth quadrature signal and thefourth quadrature local oscillator signal 674, which exerts a reverse 45degree phase shift to the fourth quadrature signal, compared to thephase shift applied by the second quadrature local oscillator signal 672to the second quadrature signal. In this manner, the fourth quadraturesignal is up-converted to a radio frequency (RF) signal by fourthquadrature local oscillator signal 674. Each of the up converted RFsignals from mixers 636 and 637 are summed within the first quadraturetransmitter path at 655.

Thereafter, signals on both first and second transmitter paths aresummed at summing junction 651. In some optional examples, thequadrature transmitter may further comprise a matched radio frequencyamplifier circuit that includes at least two sets of matched radiofrequency, RF, amplifiers, 656 each set of matched RF amplifiers 656being connected to a respective first transmitter path or secondtransmitter path. In some optional examples, the quadrature transmittermay further comprise a matched filter circuit that includes at least twosets of matched filters 644, each set of matched filters 644 beingconnected to a respective first transmitter path or second transmitterpath. In some optional examples, the quadrature transmitter may furthercomprise a matched radio frequency amplifier circuit that includes a setof matched radio frequency, RF, amplifiers, 666 located after eachtransmitter path quadrature signal has been combined, and before the twotransmitter paths are combined at summing junction 651.

It can be shown that rejection of the CIM3 and CIM5 products, as well assuppression of 3^(rd) harmonic spurs can be achieved. In some examples,the use of a single pair of DACs reduces the part count, albeit that thephase rotation has to be in an analog domain, which may introduce extranoise and may degrade SNR. In some applications, this trade-off may beadvantageous.

FIG. 7 illustrates an example overview of a still yet furtheralternative embodiment using two pairs of DACs and digital BB phaserotation (with optional error correction on each transmitter path),according to example embodiments of the invention. In the exampleembodiments, CIM3 and CIM5 are again advantageously cancelled, as wellas 3^(rd) order harmonics, which could regenerate the CIM products andfurther harmonics in the subsequent transmit chain stages, if notsupressed. Again, the transmitter architecture 700 employs a 45 degree(π/4) phase shift at the LO, and a reverse 45 degree (π/4) phase shiftat baseband (BB). In this example, the directions of the phase rotationare not specified, as it can be appreciated that the BB and LOrespectively apply reverse (i.e. opposite) 45 degree phase rotations.

The transmitter architecture 700 comprises a first quadrature (I/Q)baseband input signal 731. In this example, the first and second I/Qbaseband input signals 731, 732 may be subjected to a first (optional)batch of error correction in the digital domain in error correctioncircuit 720. Thus, in essence in this example, the error correctioncircuit 720 applies corrections to a single quadrature baseband inputsignal, of a composite transmit system implemented with two paths,before phase rotation.

The error corrected first I/Q baseband input signal is then split suchthat it also creates a first digital I/Q baseband input signal and asecond digital I/Q baseband input signal. The first and second digitalI/Q baseband input signals are routed to two separate second errorcorrection circuits 722, 724, respectively located in first and secondtransmitter paths. In the first transmitter path, after being subjectedto the separate second batch of error correction in the digital domainin error correction circuit 722, first quadrature (I/Q) baseband inputsignal 731 is then converted into the analog domain by a pair ofdigital-to-analog converters (DACs) 780. In the same manner as FIG. 4and FIG. 5, analog first I/Q baseband input signal 731 is input to firstquadrature up-mixers 734, 735, which up-convert the first I/Q basebandsignals 731 in response to respective quadrature local oscillator (LO)signals. There is a 90 degree phase shift between first quadrature LOsignal 771 and second quadrature LO signal 772. Each of the up convertedRF signals from mixers 734 and 735 are summed within the firstquadrature transmitter path at 754.

In this example, after first error correction in the digital domain inerror correction circuit 720 the single, common, digital I/Q basebandinput signal (for a composite transmit system) is input to a digitalbaseband 45 degree phase rotation circuit 790 that imparts a 45 degreephase rotation to the input signal to create a second digital I/Qbaseband input signal. Thereafter, the second digital I/Q baseband inputsignal is input to a third error correction circuit 724 and converted toanalog form in DACs 780 to produce analog second I/Q baseband inputsignals 732 for the second transmit path.

In accordance with example embodiments of the invention, the analogsecond I/Q baseband input signal 732 includes a third quadrature signalthat is a 45 degree phase shifted version of the first quadrature signalof the analog first I/Q baseband input signal 731. The analog second I/Qbaseband input signal 732 also includes a fourth quadrature signal thatis a 45 degree phase shifted version of the second quadrature signal ofthe analog first I/Q baseband input signal 731.

In some examples, the configuration to use separate error correctionblocks 722, 724 may be used to correct different imperfections in thefirst and second transmit paths, as well as any mismatch between thefirst and second transmit path. This ensures that the first and secondtransmit paths are substantially matched in order to achieve the bestpossible cancellation of CIM3/CIM5 as well as 3^(rd) order harmonicsproducts. The optional error correction 720 at the common singlebaseband input for the composite system can be used to apply furthercorrections to improve signal quality of the composite system. In oneexample correction blocks 722, 724 may correct gain and phase errors inthe first and second transmit path, whilst common correction block 720may correct the channel response (e.g. asymmetry and ripple) of acomposite system.

In the second transmitter path, after being subjected to the separatesecond batch of error correction in the digital domain in errorcorrection circuit 724, second quadrature (I/Q) baseband input signal732 is then converted into the analog domain by a pair ofdigital-to-analog converters (DACs) 780.

The second I/Q baseband input signal 732 is input to respective thirdquadrature mixer 736 and fourth quadrature mixer 737. In accordance withexample embodiments of the invention, third quadrature mixer 736receives the third quadrature signal and the third quadrature localoscillator signal 773, which exerts a reverse 45 degree phase shift tothe third quadrature signal, compared to the phase shift applied by thefirst quadrature local oscillator signal 771 to the first quadraturesignal. In this manner, the third quadrature signal is up-converted to aradio frequency (RF) signal by third quadrature local oscillator signal773. In accordance with example embodiments of the invention, fourthquadrature mixer 737 receives the fourth quadrature signal and thefourth quadrature local oscillator signal 774, which exerts a reverse 45degree phase shift to the fourth quadrature signal, compared to thephase shift applied by the second quadrature local oscillator signal 772to the second quadrature signal. In this manner, the fourth quadraturesignal is up-converted to a radio frequency (RF) signal by fourthquadrature local oscillator signal 774. Each of the up converted RFsignals from mixers 736 and 737 are summed within the first quadraturetransmitter path at 755.

Thereafter, signals on both first and second transmitter paths aresummed at summing junction 751. In some optional examples, thequadrature transmitter may further comprise a matched radio frequencyamplifier circuit that includes at least two sets of matched radiofrequency, RF, amplifiers, each set of matched RF amplifiers beingconnected to a respective first transmitter path or second transmitterpath. In some optional examples, the quadrature transmitter may furthercomprise a matched filter circuit that includes at least two sets ofmatched filters, each set of matched filters being connected to arespective first transmitter path or second transmitter path. In someoptional examples, the quadrature transmitter may further comprise amatched radio frequency amplifier circuit that includes a set of matchedradio frequency, RF, amplifiers, located after each transmitter pathquadrature signal has been combined, and before the two transmitterpaths are combined at summing junction 751.

It can be shown that rejection of the CIM3 and CIM5 products, as well assuppression of 3rd harmonic spurs can be achieved. In some examples,when using separate DACs and where digital phase rotation is employed,and where error correction circuits are separated with at least one perpath, the signal-to-noise ratio (SNR) of the transmit system may bebetter, or alternatively the noise specification may be advantageouslyrelaxed for the DACs.

FIG. 8 illustrates an example overview of a still yet furtheralternative embodiment where a sliced transmitter architecture 800 canbe selectively reconfigured to perform the harmonic/spur rejection orcancellation according to example embodiments of the invention. In thisexample, FIG. 8 introduces two additional new features, namely slicingand reconfiguration, each of which may be used in some examplearchitectures in isolation, or independently in an architecture thatuses both features.

In this example, some baseband circuitry and a large portion of the RFcircuitry is implemented on each of a plurality of slices. Notably, theplurality of slices are divided between a plurality of first transmitterpath slices 892 and a plurality of second transmitter path slices 894with the multiple outputs from both first transmitter path slices 892and second transmitter path slices 894 being combined in summingjunction 851 to form an output RF signal 850.

The sliced transmitter architecture 800 is configured such that one ormore of a number of respective transmitter slices may be selectivelyenabled, in each transmitter path. In this illustrative example, eachrespective transmitter slice 892, 894 comprises matched filters 842,quadrature up-mixer stages 834, 835, 836, 837, matched amplifiers 866and quadrature combiners 854, 855.

In the example sliced transmitter architecture 800, a yet furtheralternative transmitter architecture is illustrated that uses a 45degree (π/4) phase shift at the LO, and a reverse 45 degree (π/4) phaseshift at the quadrature baseband (BB). In this example, the directionsof the phase rotation are not specified, as it can be appreciated thatthe BB and LO respectively apply reverse (i.e. opposite) 45 degree phaserotations.

The transmitter architecture 800 comprises a first quadrature (I/Q)baseband input signal 831. In this example, the first I/Q baseband inputsignal 831 is then input to a baseband 45 degree phase rotation circuit890 that imparts a 45 degree phase rotation to the input signal tocreate an second I/Q baseband input signal 832. In the same manner asFIG. 4 and FIG. 5, first I/Q baseband input signal 831 is input to firstquadrature up-mixers 834, 835, in each first transmitter slice 892,which up-convert the first I/Q baseband signals 831 in response torespective quadrature local oscillator (LO) signals. There is a 90degree phase shift between first quadrature LO signal 871 and secondquadrature LO signal 872 that are provided by multi-phase LO generationcircuit 870. Each of the up converted RF signals from mixers 834 and 835are summed within the first quadrature transmitter path at 854.

In accordance with example embodiments of the invention, the (phaseshifted) second I/Q baseband input signal 832 includes a thirdquadrature signal that is a 45 degree phase shifted version of the firstquadrature signal of the first I/Q baseband input signal 831. The secondI/Q baseband input signal 832 also includes a fourth quadrature signalthat is a 45 degree phase shifted version of the second quadraturesignal of the first I/Q baseband input signal 831.

The second I/Q baseband input signal 832 is input to respective thirdquadrature mixer 836 and fourth quadrature mixer 837. In accordance withexample embodiments of the invention, third quadrature mixer 836receives the third quadrature signal and the third quadrature localoscillator signal 873, which exerts a reverse 45 degree phase shift tothe third quadrature signal, compared to the phase shift applied by thefirst quadrature local oscillator signal 871 to the first quadraturesignal.

In this manner, the third quadrature signal is up-converted to a radiofrequency (RF) signal by a third quadrature local oscillator signal 873.In accordance with example embodiments of the invention, fourthquadrature mixer 837 receives the fourth quadrature signal and thefourth quadrature local oscillator signal 874, which exerts a reverse 45degree phase shift to the fourth quadrature signal, compared to thephase shift applied by the second quadrature local oscillator signal 872to the second quadrature signal. In this manner, the fourth quadraturesignal is up-converted to a radio frequency (RF) signal by fourthquadrature local oscillator signal 874. Each of the up converted RFsignals from mixers 836 and 837 are summed within the first quadraturetransmitter path at 855.

Thereafter, transmitter signals on both first and second transmitterpaths, selected from one or multiple slices, are summed at summingjunction 851. In some optional examples, the quadrature transmitter mayfurther comprise a matched radio frequency amplifier circuit thatincludes at least two sets of matched radio frequency, RF, amplifiers,each set of matched RF amplifiers being connected to a respective firsttransmitter path or second transmitter path. In some optional examples,the quadrature transmitter may further comprise a matched filter circuit842 on each slice that includes at least one set of matched filters,each set of matched filters being located on a respective firsttransmitter path or second transmitter path. In some optional examples,the quadrature transmitter may further comprise a matched radiofrequency amplifier circuit that includes a set of matched radiofrequency, RF, amplifiers, 866 located on each slice before (as shown)or after each transmitter path quadrature signal has been combined, andbefore the two transmitter paths are combined at summing junction 851.

In accordance with the sliced transmitter architecture of FIG. 8, in oneexample, the sliced architecture may benefit from a controller 814 thatis able to switch on/off various sliced paths and various phaserotations options, if it is/they are not needed. By switching off thebaseband 45 degree phase shift rotation circuit 890 and the use of thereverse 45 degree phase shift rotation imparted by multi-phase LOgeneration circuit 870, the transmitter may be selectively reconfiguredto perform as a conventional transmitter.

In some examples, it is recognised that spurious emissions aredeterministic, and thus controller 814 may be arranged to turncancellation on/off based on known band information.

It can be shown that harmonic rejection of the CIM3 and CIM5 products,as well as suppression of 3^(rd) harmonic spurs can be achieved. Table 1illustrates the improvements provided by the example embodimentsdescribed herein, as compared to the known art of Ingels.

TABLE 1 Prior art of Examples described herein Ingels LO using LO ± 45deg. + BB ± 45 deg. BB ± 45 deg. LO phase (single ended) 2 4 BB phase(single ended) 4 4 CIM3 Cancel Cancel CIM5 Remain Cancel 2flo − 2fbbRemain Generated CIM3 cancelled 3flo − fbb Remain Cancel 3flo + 7fbbRemain Cancel Power Loss Significant No loss

In some examples, the use of slicing reduces overall power consumptionand provides increased flexibility and programmability, as respectiveslices can be enabled/disabled. In some applications, this trade-off ofincreased part count (due to multiple slices) may be advantageous.

Although FIG. 8 is illustrated with regard to slicing only for theup-mixer and programmable gain amplifier re-configurability, it isenvisaged that in other examples the slices may be extended to includesets of DACs and in some examples the analog quadrature basebandcomponents.

FIG. 9 illustrates a graphical example 900 of a simulation of phasesensitivity, highlighting the attenuation achieved for respectivemultiple harmonics or spurs, according to example embodiments of theinvention. The X-axis 910 shows the phase shift applied to thequadrature baseband input and LO input (reverse shift) of the secondtransmitter path with respect to the quadrature baseband input and LOinput of the first transmitter path. The Y-axis 905 shows the level ofundesired CIM3, CIM5 and 3^(rd) harmonic spurs at the output of thetransmitter system relative to the desired output signal in dBc. With arelative phase shift of zero degrees between the first transmitter pathand second transmitter path the system essentially reverts to atraditional quadrature transmitter. Hence points on the right hand sideof FIG. 9 at zero degrees can be interpreted as the performance of atraditional transmit architecture. As described in detail for severalexample embodiments in the earlier description, FIG. 9 clearlydemonstrates that significant rejection of CIM3, CIM5 and 3^(rd)harmonic products is achieved for a relative phase shift of 45 degrees.For instance, the CIM3 product represented by line 922 shows aperformance of −50 dBc for a traditional transmit architecture withoutphase shift but improves by 40 dB to −90 dBc when 45 degree phase shiftis applied. Similarly, the CIM5 product 932, the (3fo−fbb) product 912and the (3fo+7fbb) product 942 are all attenuated by 35-40 dB when 45degree phase shift is applied.

As illustrated, the spur and harmonic cancellation is mathematicallyideal for a phase shift of exactly 45 degrees, if the two transmit pathsare ideally matched. In reality, however, perfect cancellation may notbe achievable because the two transmit paths will always have somemismatch and indeed the 45 degree phase rotation may have some errortoo, with FIG. 9 showing the sensitivity of the achievable cancellationto errors in the phase rotation. Although the best cancellation isachieved for an ideal 45 degree phase shift, it has been found that with+/−1 degree errors, the system still achieves very good cancellation(within a few dB of the optimum). Even with phase shift errors of +/−5degrees it has been found that the system provides good, meaningfulcancellation of around 20 dB for CIM3, CIM5 and 3rd harmonic. Thus,examples of the invention, and thus the claims, are targeted for anysystem where the phase shift is substantially 45 degrees+/−5 degrees, toallow for component tolerances, matched path differences and associatederror.

FIG. 10 illustrates a graphical example 1000 of a simulation ofamplitude sensitivity highlighting the attenuation achieved forrespective multiple harmonics or spurs, according to example embodimentsof the invention. The relative normalised spur level on the Y-axis 1005,of the same four spurs 1012, 1022, 1032, 1042 as in FIG. 9, is shown. Inthis case, the X-axis 1010 is the gain mismatch between the firsttransmitter path and the second transmitter path. FIG. 10 clearly showsthat good cancellation is achieved when the gain of the first and secondtransmit paths are essentially matched. Furthermore, FIG. 10 highlightsthat good spur rejection can be achieved without very high gainaccuracy, and even with a large +/−2 dB gain mismatch between the firstand second transmitter path the cancellation is degraded by only 10-15dB.

Referring now to FIG. 11, an example flowchart 1100 illustrates a methodto suppress, cancel or reject multiple harmonics or spurs, according toexample embodiments of the invention. The flowchart 1100 is describedfor a transmitter that comprises a first transmitter path and a secondtransmitter path that are matched. The method comprises: receiving afirst sets of quadrature signals at the first transmitter path at 1102;and receiving a second sets of quadrature signals at the secondtransmitter path at 1104. The first set of quadrature signals is a ±45°phase shifted version of the second set of quadrature signals; and thefirst set of quadrature LO signals is a reverse ±45° phase shiftedversion of the second set of quadrature LO signals. The flowchart thenincludes generating respective first and second sets of quadrature LOsignals at 1106. In some optional examples, such as that described inFIG. 8, a determination as to whether spur suppression may be needed maybe made at 1108, for example as determined by a spurious emissioncircuit.

If the determination at 1108 is that no spur suppression may be needed,a controller may decide to not turn off the second transmitter path, butre-configure the second transmitter path to use the same baseband and LOphases as the first transmitter path. However, if the determination at1108 is that spur suppression may be needed, the method furthercomprises multiplying the first and second sets of quadrature signalswith the respective first or second sets of quadrature LO signals toproduce a respective output radio frequency, RF, signal at 1110.Thereafter, the multiple outputs of the first transmitter path and thesecond transmitter path are combined to produce an output radiofrequency signal at 1112.

The previous examples in FIGS. 3-11 have described error correctiontechniques whereby the error correction is performed purely at basebandfrequencies, i.e. before any up-conversion of the quadrature signal to aradio frequency or intermediate frequency, based on the LO frequency.Mathematically, looking closely at the CIM3 product for two differenttransmit frequencies (Tx1 and Tx2) at a frequency: flo−3fbb, the CIM3product can be described by the equations [4] and [5] as follows:

TX1: A sin(2H(f _(L)−3f _(B))+φ_(L,1)−3φ_(B,1))  [4]

and TX2: A sin(2H(f _(L)−3f _(B))+φ_(L,2)−3φ_(B,2))  [5]

Clearly, from [4] and [5], the phase difference between the two paths is180° as can be observed from: Δφ_(L)−3Δφ_(B), if Δφ_(L)=±45°,Δω_(L)=∓45°. Thus, in a practical implementation and due to any circuitmismatches, phase differences at both LO and BB contain errors from theideal value.

When phase error correction is performed at BB, the phase difference canbe defined as:

±180°+φ_(E,L)−3φ_(E,B)+φ_(c)  [6]

and with φ_(c)=3φ_(E,B)−φ_(E,L), CIM3 cancellation degradation due tophase difference is recovered

However, if the same calculations are performed at the 3flo−fbb spur,the resultant phase difference can be defined as:

±180°+3φ_(E,L)−φ_(E,B)+φ_(c)  [7]

Hence, if the correction for CIM3 is applied, a resulting 3flo−fbb phaseerror exists, as illustrated in equation [8]:

±180°+3φ_(E,L)−φ_(E,B)+φ_(c)=±180°+2φ_(E,L)+2φ_(E,B)  [8]

Thus, it may be observed that the baseband phase error correction cancorrect only one spur cancellation at any instant in time.

Therefore further examples of the invention propose a solution wherebyLO phase error is corrected at LO frequencies and BB phase error iscorrected at BB frequencies.

Referring now to FIG. 12, an example overview of a still yet furtheralternative embodiment where a sliced transmitter architecture 1200 canbe selectively reconfigured to correct for LO phase error at LOfrequencies and correct for BB phase error at BB frequencies, accordingto example embodiments of the invention. In this example, FIG. 12introduces a further additional new feature, namely correcting LO phaseerror at an LO frequency and correcting baseband phase error at abaseband frequency, for example within a slicing and reconfigurationarchitecture. It is envisaged that the approach of FIG. 12 may also beused in any of the previous example architectures, and is not limited tothe specific architecture illustrated. For example, it is envisaged thatthe BB phase error correction can also be located before the DACs, asillustrated in FIG. 6 and FIG. 7.

The transmitter architecture 1200 includes first and second I/Q basebandinput signals that are converted from a digital form to an analog formby a pair of DACs 1280 to produce a first quadrature (I/Q) basebandinput analog signal 1231. Thus, in essence in this example, the BB phaseerror correction circuit 1290 applies corrections to a single I/Qbaseband input signal of a composite transmit system implemented withtwo paths.

In this example, some baseband circuitry and a large portion of the RFcircuitry is implemented on each of a plurality of slices. Notably, theplurality of slices are divided between a plurality of first transmitterpath slices 1292 and a plurality of second transmitter path slices 1294with the multiple radio frequency outputs from both first transmitterpath slices 1292 and second transmitter path slices 1294 being combinedin summing junction 1251 to form an output RF signal 1250.

The sliced transmitter architecture 1200 is configured such that one ormore of a number of respective transmitter slices may be selectivelyenabled, in each transmitter path. In this illustrative example, eachrespective transmitter slice 1292, 1294 comprises matched filters 1242,quadrature up-mixer stages 1234, 1235, 1236, 1237, matched amplifiers1266 and quadrature combiners 1254, 1255.

The first quadrature (I/Q) baseband input analog signal 1231 is inputdirect to first transmitter path slices 1292 and either direct to secondtransmitter path slices 1294 or to second transmitter path slices 1294via a rotation and correction circuit, dependent upon the configurationof controllable switches 1246.

In this example, the first I/Q baseband input signal 1231 is then inputto a baseband 45 degree phase rotation circuit 1290 that imparts a 45degree phase rotation to the input signal to create a second I/Qbaseband 45 degree phase rotated and phase error corrected input signal1232. In the same manner as FIG. 4 and FIG. 5, first I/Q baseband inputsignal 1231 is input to first quadrature up-mixers 1234, 1235, in eachfirst transmitter slice 1292, which up-convert the first I/Q basebandsignals 1231 in response to respective quadrature local oscillator (LO)signals. Each of the up converted RF signals from mixers 1234 and 1235are summed within the first quadrature transmitter path at 1254.

In accordance with example embodiments of the invention, the (phaseshifted and phase error corrected) and second I/Q baseband 45 degreephase rotated and phase error corrected input signal 1232 includes athird quadrature signal that is a 45 degree phase shifted version of thefirst quadrature signal of the first I/Q baseband input signal 1231. Thesecond I/Q baseband 45 degree phase rotated and phase error correctedinput signal 1232 also includes a fourth quadrature signal that is a 45degree phase shifted version of the second quadrature signal of thefirst I/Q baseband input signal 1231.

In the example sliced transmitter architecture 1200, a 45 degree (π/4)phase shift is performed at the LO frequency, and a reverse 45 degree(π/4) phase shift is performed at the quadrature baseband (BB). Notably,in this example, the 45 degree (π/4) phase shift at BB includes a BBphase error correction configured to correct for a BB phase error:45°+φ_(E,B), which is implemented in baseband 45 degree phase rotationand BB phase error correction circuit 1290.

There is a 90 degree phase shift between first quadrature LO signal 1271and second quadrature LO signal 1272 that are provided by multi-phase LOgeneration and correction circuit 1270. Furthermore, in this example,the LO frequency includes a LO phase error correction to correct for aLO phase error, which is implemented in multi-phase generation andcorrection circuit 1270 that includes LO phase error correction at180°±45° phase rotation values, i.e. 45°+φ_(E,L) and 135°+φ_(E,L).

In this example, the directions of the phase rotation are not specified,as it can be appreciated that the BB and LO respectively apply reverse(i.e. opposite) 45 degree phase rotations.

The second I/Q baseband 45 degree phase rotated and phase errorcorrected input signal 1232 is input to respective third quadraturemixer 1236 and fourth quadrature mixer 1237. In accordance with exampleembodiments of the invention, third quadrature mixer 1236 receives thethird quadrature signal and the third quadrature local oscillator signal1273, which exerts a reverse 45 degree phase shift plus phase errorcorrected input to the third quadrature signal, compared to the phaseshift applied by the first quadrature local oscillator signal 1271 tothe first quadrature signal.

In this manner, the third quadrature signal is up-converted to a radiofrequency (RF) signal by a third quadrature local oscillator signal1273. In accordance with example embodiments of the invention, fourthquadrature mixer 1237 receives the fourth quadrature signal and thefourth quadrature local oscillator signal 1274, which exerts a reverse45 degree phase shift plus phase error corrected input to the fourthquadrature signal, compared to the phase shift applied by the secondquadrature local oscillator signal 1272 to the second quadrature signal.In this manner, the fourth quadrature signal is up-converted to a radiofrequency (RF) signal by fourth quadrature local oscillator signal 1274.Each of the up converted RF signals from mixers 1236 and 1237 are summedwithin the first quadrature transmitter path at 1255.

Thereafter, transmitter signals on both first and second transmitterpaths, selected from one or multiple slices, are summed at summingjunction 1251. In some optional examples, the quadrature transmitter mayfurther comprise a matched radio frequency amplifier circuit thatincludes at least two sets of matched radio frequency, RF, amplifiers,each set of matched RF amplifiers being connected to a respective firsttransmitter path or second transmitter path. In some optional examples,the quadrature transmitter may further comprise a matched filter circuit1242 on each slice that includes at least one set of matched filters,each set of matched filters being located on a respective firsttransmitter path or second transmitter path. In some optional examples,the quadrature transmitter may further comprise a matched radiofrequency amplifier circuit that includes a set of matched radiofrequency, RF, amplifiers, 1266 located on each slice before (as shown)or after each transmitter path quadrature signal has been combined, andbefore the two transmitter paths are combined at summing junction 1251.

In accordance with the sliced transmitter architecture of FIG. 12, inone example, the sliced architecture may benefit from a controller 1214that is able to switch on/off various sliced paths and various phaserotation options, if it is/they are not needed. By switching off thebaseband 45 degree phase rotation and BB phase error correction circuit1290 and the use of the reverse 45 degree phase shift rotation and LOphase error correction imparted by multi-phase LO generation andcorrection circuit 1270, the transmitter may be selectively reconfiguredto perform as a conventional transmitter.

In some examples, it is recognised that spurious emissions aredeterministic, and thus controller 1214 may be arranged to turncancellation on/off based on known band information.

In some examples, it is envisaged that the phase error correctionarchitecture may comprise three error correction circuits operablycoupled to an output of the set of DACs wherein two BB error correctioncircuits 1290, 1291 are configured to correct non-idealities (e.g. BBphase errors) within a respective first and second transmitter path anda third error correction circuit is configured to correct non-idealities(e.g. LO phase errors) on both the first and second transmitter paths.

In some examples, the use of slicing reduces overall power consumptionand provides increased flexibility and programmability, as respectiveslices can be enabled/disabled. In some applications, this trade-off ofincreased part count (due to multiple slices) may be advantageous.

FIG. 12 illustrates a further exemplary transmitter architecture 1201,whereby one or more transmitter amplifier(s), e.g. a power amplifier1296 or an amplification chain or a part of an amplifier chain, may belocated outside of the first transmitter path slices 1292 and secondtransmitter path slices 1294. Here, in this example, it is envisagedthat the signal combiner of the first transmitter path and secondtransmitter path may reside before one or more or all of the RF signalamplification component(s), as illustrated with power amplifier 1296being located after summing junction 1251 that sums the RF signals inboth the first transmitter path and second transmitter path. In thisexample, a reduction in a number of RF signal amplification component(s)may be achieved. In this example, lesser consideration needs to be givenby a designer to the combination of multiple higher power RF signals bysumming junction 1251.

Although FIG. 12 is illustrated with regard to slicing only for theup-mixer and programmable gain amplifier re-configurability, it isenvisaged that in other examples the slices may be extended to includesets of DACs and, in some examples, the analog quadrature basebandcomponents.

Referring now to FIG. 13, an example flowchart 1300 illustrates a methodto suppress, cancel or reject multiple harmonics or spurs with bothbaseband phase rotation and local oscillator phase rotation, accordingto example embodiments of the invention. The flowchart 1300 is describedfor a transmitter that comprises a first transmitter path and a secondtransmitter path that are matched. The method comprises: receiving afirst sets of quadrature signals at the first transmitter path at 1302;and receiving a second sets of quadrature signals at the secondtransmitter path at 1304. The first set of quadrature signals is a ±45°phase shifted version of the second set of quadrature signals; and thefirst set of quadrature LO signals is a reverse ±45° phase shiftedversion of the second set of quadrature LO signals. The flowchart thenincludes generating respective first and second sets of quadrature LOsignals at 1306. In some optional examples, such as that described inFIG. 8, a determination as to whether spur suppression may be needed maybe made at 1308, for example as determined by a spurious emissioncircuit.

If the determination at 1308 is that no spur suppression may be needed,at 1309, a controller may decide to not turn off the second transmitterpath, but re-configure the second transmitter path to use the samebaseband and LO phases as the first transmitter path. However, if thedetermination at 1308 is that spur suppression may be needed, the methodfurther includes, at 1310 and according to the methodology adopted anddescribed with respect to FIG. 12, BB phase error is corrected at BBfrequencies. At 1312, a determination is made as to whether multiplespur cancellations are needed.

If, at 1312, multiple spur cancellation are not needed, the flowchartskips to 1316. However, if at 1312, multiple spur cancellations areneeded, and according to the methodology adopted as described withrespect to FIG. 12, LO phase error is corrected at LO frequencies at1314.

Thereafter, the first and second sets of quadrature signals aremultiplied with the respective first or second sets of quadrature LOsignals in order to produce a respective output radio frequency, RF,signal at 1316. Thereafter, the multiple outputs of the firsttransmitter path and the second transmitter path are combined to producean output radio frequency signal at 1318.

It is envisaged that the aforementioned inventive concept can be appliedby a semiconductor manufacturer to any radio frequency transmittermodule comprising baseband and/or radio frequency components or circuitsthat supports quadrature signals. It is further envisaged that, forexample, a semiconductor manufacturer may employ the inventive conceptin a design of a stand-alone radio frequency transmitter module orapplication-specific integrated circuit (ASIC) or may implement theconcepts herein described in any other sub-system element.

However, it will be appreciated by a skilled artisan that the inventiveconcept herein described may be embodied in any type of wirelesscommunication unit, such as those used in mobile phone communications,radar applications and/or military, civil and land mobile radioapplications, to name but a few potential applications. In someexamples, die may be constructed using one or more of the followingtechnologies: complementary metal-oxide semiconductor (CMOS), BiCMOS(where BiCMOS is an evolved semiconductor technology that integrates twoformerly separate semiconductor technologies, those of the bipolarjunction transistor and the CMOS transistor, in a single integratedcircuit device) or gallium arsenide (GaAs).

It will be appreciated that any suitable distribution of functionalitybetween different functional units, for example with respect to theintegrated circuits, may be used without detracting from the invention.Hence, references to specific functional units are only to be seen asreferences to suitable means for providing the described functionality,rather than indicative of a strict logical or physical structure ororganization.

Although the present invention has been described in connection withsome embodiments, it is not intended to be limited to the specific formset forth herein. Rather, the scope of the present invention is limitedonly by the accompanying claims. Additionally, although a feature mayappear to be described in connection with particular embodiments, oneskilled in the art would recognize that various features of thedescribed embodiments may be combined in accordance with the invention.In the claims, the term ‘comprising’ does not exclude the presence ofother elements or steps.

Furthermore, although individually listed, a plurality of means,elements or method steps may be implemented by, for example, a singleunit or processor. Additionally, although individual features may beincluded in different claims, these may possibly be advantageouslycombined, and the inclusion in different claims does not imply that acombination of features is not feasible and/or advantageous. Also, theinclusion of a feature in one category of claims does not imply alimitation to this category, but rather indicates that the feature isequally applicable to other claim categories, as appropriate.

Furthermore, the order of features in the claims does not imply anyspecific order in which the features must be performed and in particularthe order of individual steps in a method claim does not imply that thesteps must be performed in this order. Rather, the steps may beperformed in any suitable order. In addition, singular references do notexclude a plurality. Thus, references to ‘a’, ‘an’, ‘first’, ‘second’,etc. do not preclude a plurality.

Thus, an improved transmitter and method for reducing or cancellingharmonic spurs, including CIM products has been described, wherein theaforementioned disadvantages with prior art arrangements have beensubstantially alleviated.

We claim:
 1. A quadrature transmitter comprising: a first transmitter path and a second transmitter path that are matched and wherein each transmitter path comprises: at least one input arranged to receive respective first or second sets of quadrature baseband signals; at least one local oscillator, LO, port configured to receive respective first and second sets of quadrature LO signals; and at least one mixer stage coupled to the at least one input and configured to respectively multiply the sets of quadrature baseband signals with the respective first or second sets of quadrature LO signals to produce a respective output radio frequency, RF, signal; and a combiner configured to combine the output radio frequency signals of the first transmitter path and the second transmitter path; wherein: the first set of quadrature baseband signals is a substantially 45° phase shifted version of the second set of quadrature baseband signals; and the first set of quadrature LO signals is a reverse substantially 45° phase shifted version of the second set of quadrature LO signals; and wherein at least one of the first transmitter path and second transmitter path comprises a baseband error correction circuit configured to correct a phase error between the first or second sets of quadrature baseband signals at baseband and a LO error correction circuit configured to correct a phase error between the first or second sets of quadrature baseband signals at a LO frequency.
 2. The quadrature transmitter of claim 1 wherein at least two sets of undesired radio frequency signals are anti-phase such that they are cancelled in the combiner.
 3. The quadrature transmitter of claim 1 wherein a single set of quadrature baseband signals is applied to both the first transmitter path and the second transmitter path and wherein only one of the first and second transmitter paths comprises at least one baseband phase shifter coupled to the quadrature input and configured to provide a phase shifted by substantially ±45° representation of the quadrature input signal applied to the other transmitter path to its respective baseband input.
 4. The quadrature transmitter of claim 1 further comprising a pair of digital to analog converters, DACs, configured to generate analog quadrature signals wherein the baseband error correction circuit is operably coupled to an output of the pair of DACs at thereby effecting baseband phase error correction in an analog domain.
 5. The quadrature transmitter of claim 4 wherein the baseband phase error correction in an analog domain is performed after a phase rotation of a set of quadrature analog signals.
 6. The quadrature transmitter of claim 4 further comprising three error correction circuits operably coupled to an input of the set of DACs wherein two error correction circuits are configured to correct non-idealities within a respective first and second transmitter path and a third error correction circuit is configured to correct non-idealities on both the first and second transmitter paths.
 7. The quadrature transmitter of claim 4 further comprising one baseband error correction circuit operably coupled to an output of the set of DACs configured to apply one or more phase error correction to the set of quadrature digital signals shared between the first and second transmitter paths.
 8. The quadrature transmitter of claim 1 further comprising a controller coupled to first pairs of switches coupled to the baseband input of the second transmit path and second pairs of switches located on the LO path of the second transmitter path and configured to selectively reconfigure the second transmitter path to operate on the same set of LO and baseband signals as the first transmitter path.
 9. The quadrature transmitter of claim 1 wherein each of the first transmitter path and second transmitter path is implemented as a plurality of sliced transmitter paths wherein the combiner is a power combiner located external to the first plurality of sliced RF modules and second plurality of sliced RF modules.
 10. The quadrature transmitter of claim 9 wherein each of the first plurality of sliced RF modules and second plurality of sliced RF modules comprises a second combiner configured to combine RF quadrature signals output by respective sliced RF modules.
 11. The quadrature transmitter of claim 10 further comprising a controller coupled to: a first pair of switches coupled to the baseband input of the second transmit path to provide the first set of quadrature baseband signals to each sliced RF module of the second transmitter path; and second pairs of switches located on the LO path of each respective sliced RF module of the second transmitter path and configured to selectively apply the reverse phase shifts to mixer stages on each sliced RF module of the second transmitter path.
 12. The quadrature transmitter of claim 1 further comprising at least two sets of matched radio frequency, RF, amplifiers, each set of matched RF amplifiers being connected to a respective first transmitter path or second transmitter path.
 13. The quadrature transmitter of claim 1 wherein the combiner configured to combine the output radio frequency signals of the first transmitter path and the second transmitter path is located before at least one transmitter amplifier in a transmitter chain.
 14. A communication unit comprising a quadrature transmitter that comprises: a first transmitter path and a second transmitter path that are matched and wherein each transmitter path comprises: at least one input arranged to receive respective first or second sets of quadrature baseband signals; at least one local oscillator, LO, port configured to receive respective first and second sets of quadrature LO signals; and at least one mixer stage coupled to the at least one input and configured to respectively multiply the sets of quadrature baseband signals with the respective first or second sets of quadrature LO signals to produce a respective output radio frequency, RF, signal; and a combiner configured to combine the output radio frequency signals of the first transmitter path and the second transmitter path; wherein: the first set of quadrature baseband signals is a substantially 45° phase shifted version of the second set of quadrature baseband signals; and the first set of quadrature LO signals is a reverse substantially 45° phase shifted version of the second set of quadrature LO signals; and wherein at least one of the first transmitter path and second transmitter path comprises a baseband error correction circuit configured to correct a phase error between the first or second sets of quadrature baseband signals at baseband and a LO error correction circuit configured to correct a phase error between the first or second sets of quadrature baseband signals at a LO frequency.
 15. The communication unit of claim 14 wherein at least two sets of undesired radio frequency signals are anti-phase such that they are cancelled in the combiner.
 16. The communication unit of claim 14 further comprising a pair of digital to analog converters, DACs, configured to generate analog quadrature signals wherein the baseband error correction circuit is operably coupled to an output of the pair of DACs at thereby effecting baseband phase error correction in an analog domain.
 17. The communication unit of claim 16 further comprising three error correction circuits operably coupled to an input of the set of DACs wherein two error correction circuits are configured to correct non-idealities within a respective first and second transmitter path and a third error correction circuit is configured to correct non-idealities on both the first and second transmitter paths.
 18. The communication unit of claim 16 further comprising one baseband error correction circuit operably coupled to an output of the set of DACs configured to apply one or more phase error correction to the set of quadrature digital signals shared between the first and second transmitter paths.
 19. The communication unit of claim 14 further comprising a controller coupled to first pairs of switches coupled to the baseband input of the second transmit path and second pairs of switches located on the LO path of the second transmitter path and configured to selectively reconfigure the second transmitter path to operate on the same set of LO and baseband signals as the first transmitter path.
 20. A method for a transmitter that comprises a first transmitter path and a second transmitter path that are matched, the method comprising: receiving first sets of quadrature baseband signals at the first transmitter path; receiving second sets of quadrature baseband signals at the second transmitter path; correcting a phase error between the first or second sets of quadrature baseband signals at baseband by a baseband error correction circuit; generating respective first and second sets of quadrature LO signals; correcting a phase error between the first or second sets of quadrature baseband signals at a LO frequency by a LO error correction circuit; multiplying the first and second sets of quadrature baseband signals with the respective first or second sets of quadrature LO signals to produce a respective output radio frequency, RF, signal; and combining the output radio frequency signals of the first transmitter path and the second transmitter path; wherein: the first set of quadrature baseband signals is a substantially 45° phase shifted version of the second set of quadrature baseband signals; and the first set of quadrature LO signals is a reverse substantially 45° phase shifted version of the second set of quadrature LO signals. 